Waveguide to microstrip transition

ABSTRACT

A waveguide coupling device comprises a waveguide section in which a guided wave may be propagated in at least one waveguide mode and which has two slits in one of its walls. The waveguide mode has a field component parallel to the slotted wall with a nodal plane oriented in the longitudinal direction of the waveguide section, and/or it induces in the walls of the waveguide section a wall current distribution with just such a nodal plane. The slits lie on opposing sides of the nodal plane. One or two antenna sections bridge the first slit or both slits.

The present invention concerns a device for coupling a radio frequencysignal propagating in a metallic conductor into a waveguide or from awaveguide into a metallic conductor.

Conventional coupling devices of this type comprise a waveguide sectionin which a guided wave is capable of propagating in at least onewaveguide mode and which has a slit in one of its walls, through whichthe field of the waveguide mode emerges and is capable of exciting anoscillation in an antenna section arranged outside the waveguidesection, bridging the slit.

Only a part of the radio frequency energy emerging through the slit isactually utilised for exciting the oscillation in the antenna section;the remainder is radiated into the free space lying above the slit. Thisis undesirable, not only because the energy is thereby radiated unused,but because it may have an interfering influence on equipment componentssituated in the free space.

If, for instance, a coupling device of this type is used in a groupantenna in order, via slits in the walls of a waveguide and antennasections arranged crossing it, to feed individual antenna elements ofthe group antenna, then the interference radiation emerging from theslits may sensitively impair the field pattern of the group antenna.

In W. Keusgen and B. Rembold, “broadband Planar Subarray for MicrowaveWLAN Applications”, MIOP, Stuttgart, 2001, it is proposed to circumventthis problem in that the interference radiation is coupled into aradiator element which actively contributes to the function of the groupantenna. This solution involves a significant calculation effort,however, and is not generally applicable.

In F. J. Villegas, D. I. Stones, H. A. Hung: “A NovelWaveguide-to-Microstrip Transition for Millimetre Wave Applications”,IEEE Trans. on Microwave Theory and Techniques, vol. 47, No. 1, January1999, it is proposed that the interference radiation be suppressed withthe aid of cover caps placed over the respective slits to preventemergence of the interference radiation. However, this solution iscomplex in the implementation, since for every slit such a cover withfed-through antenna section is required.

It is an aim of the present invention to provide a waveguide couplingdevice of the aforementioned type in which the emergence of interferenceradiation is effectively suppressed in simple manner and which may bemanufactured with little effort.

This aim is fulfilled by providing, in the side wall which has the firstslit, a second slit which is so arranged that the two slits lie onopposite sides of a nodal line of a field component of the waveguidemode that is oriented parallel to the slotted wall.

The invention is preferably applied to a waveguide of rectangularcross-section and particularly to its principal mode, known as themagnetic fundamental wave or the H₁₀ wave. Based on the explanationsgiven here, however, a person skilled in the art will be able to applythe invention also to other waveguide cross-sections and waveguidemodes.

If a coordinate system is established whereby the X-axis isperpendicular to a narrow side wall and the Y-axis is perpendicular to abroad side wall of the waveguide section and the Z-axis extends in thelongitudinal direction of the waveguide section, the H₁₀ wave has fieldcomponents H_(x) and H_(z) parallel to a broad side wall of thewaveguide. Of these components, the component H_(z) has a nodal plane,which extends in the longitudinal direction of the waveguide section andintersects its two broad side walls centrally. The H_(z) component hasopposite signs on the different sides of the nodal plane. Thus thefields emerging from the two slits and originating from the H_(z)component oscillate with opposing phase and tend to cancel each otherout in the radiation zone. The E_(y) component of the H₁₀ wave excites,in the side walls of the waveguide section, cross-currents which flow inopposing directions on either side of the same nodal plane and evokeopposite-oriented electric fields in the X-direction at the two slits.These also tend to cancel each other out in the radiation zone.

This cancellation is all the more complete, the more symmetrical is thearrangement of the two slits in relation to the nodal plane. If thelocus of one slit is the reflection of the other with respect to thenodal plane, then their E_(x) components compensate each othercompletely in the radiation zone on the nodal plane, provided thesymmetry is not broken by the antenna section crossing the first slit,and are severely reduced laterally by this compared with the field of awaveguide section having a single slit.

With an inversion symmetry arrangement of the slits relative to a pointin the nodal plane, that is to say, the locus of one slit is theinverted reflection of the other with respect to the nodal plane,sufficient compensation may also be achieved, provided the extent of theslits in the Z-direction is significantly smaller than the wavelength ofthe waveguide mode and thus phase differences between the fields atinversion symmetrical points of the two slits can be ignored.

The antenna section is in general linked at one end to a conductor forconducting away the coupled-in RF signal and free at its other end. Thisfree end may preferably be placed at a distance of λ_(s)/4 from theslit, either fixed or adjustable, where λ_(s) is the wavelength of thesignal induced in the antenna section. This achieves the result that aportion of the coupled-in signal propagating in the antenna section fromthe slit directly in the direction of the connecting conductor and aportion initially reflected at the free end are constructively combined,so that a strong coupling is achieved.

In order to avoid break of symmetry through an intersecting antennasection, a second antenna section may advantageously be arrangedbridging the second slit. This antenna section may be employed forfeeding a different RF component from that fed by the first antennasection, or for feeding the same RF component.

According to a preferred embodiment, in the latter case, two antennasections are linked at one point parallel to a connecting conductor,i.e. they each have one end linked to the connecting conductor and onefree end.

The antenna sections may be so arranged that they cross the slitsassigned to them in respective opposing directions, i.e. their free endseither both lie between the slits or both beyond the slits. In this caseit is preferred that the antenna sections should have a total length Lbetween (n−⅜)λ_(s) and (n+⅜)λ_(s), where n is an integer and λ_(s) isthe wavelength of the oscillation induced in the antenna sections by theguided wave. If L is exactly equal to nλ_(s), then the oscillationscoupled at the two slits in the antenna sections interfere exactlycophasally and optimum coupling is achieved. Values deviating fromnλ_(s) may be used if a weaker coupling is desired.

If, on the other hand, the antenna sections cross their slits in thesame direction, i.e. if the free end of one antenna section lies betweenthe slits and that of the other lies beyond the slits, then theoscillations induced at the slits interfere cophasally at a total lengthL of (n+½)λ_(s), by reason of which a total length L of between(n+⅛)λ_(s) and (n+⅞)λ_(s) is preferred.

Another possibility is to link the two antenna sections in series; inthis case, for a cophasal superposition of the oscillations induced atthe two slits, a spacing between the slits measured along the antennasections of approximately nλ_(s) if the antenna sections cross the slitsin opposing directions, or of approximately (n+½)λ_(s) is required ifthe antenna sections cross the slits in the same direction.

Preferably, the crossing points of the antenna sections with the slitslie on a line perpendicular to the longitudinal direction of thewaveguide section or to the nodal plane.

It is thereby ensured that the two antenna sections are exposed tocophasal exciting fields emerging from the slits, independently of theexact position in which the antenna sections are arranged in relation tothe waveguide section. It is particularly suitable if the antennasections lie, at least in the region of the crossing points, on a commonline, so that the phase coincidence of the fields to which the twoantenna sections are exposed is maintained even on transversedisplacement of the antenna sections.

According to a first preferred embodiment, the two slits are parallel toeach other and to the nodal plane, so that the coupling strength doesnot depend on the position of the antenna sections in the propagationdirection of the guided wave (the Z-direction), but is determinedexclusively by the position of the antenna sections transverse to thenodal plane, i.e. by the spacing of their crossing points from the freeends.

According to a second preferred embodiment, the slits run parallel andinclined to the nodal plane. The degree of deviation from parallelisminfluences the strength of the H_(z) field emerging from the slits andcoupling into the antenna sections and thus the coupling constant of thecoupling device. In particular if the slits are arranged on a rotatablewall section of the waveguide section, by rotation of this wall section,the coupling constant may be adjusted as required.

According to a third preferred embodiment, the slits have a spacingvarying along the nodal plane and the antenna sections are positionablein different positions along the nodal plane. In this case, the couplingcoefficient may be set by suitable positioning of the antenna sectionsalong the nodal plane. The nearer the slits lie to the nodal plane, thesmaller is the field component parallel to the wall in the waveguidebehind the slits and the smaller are the wall currents induced at thesite of the slits, and the smaller therefore is the emerging field towhich the antenna sections are exposed.

In the first and third embodiments, it may be provided that, whenmanufacturing the coupling device, the antenna sections are firmlyplaced at a site, whereby the antenna sections may be fixed at severalpositions on the waveguide section and the position in an individualcase is selected on the basis of a desired coupling coefficient.Alternatively, the possibility exists of providing a device foradjusting the antenna sections relative to the slits, in order also tobe able to adapt the coupling coefficients of the finished couplingdevice to requirements at any time.

Further features and advantages of the invention are given in thefollowing description of examples by reference to the attached drawings,in which

FIG. 1 shows a perspective view of a coupling device according to afirst embodiment of the invention;

FIG. 2 shows the distribution of the cross-currents in the wall of thewaveguide section of the coupling device according to FIG. 1;

FIG. 3 shows a second embodiment of a coupling device according to theinvention in a perspective view analogous to FIG. 1;

FIG. 4 shows an instantaneous current and voltage distribution in theantenna sections and the connecting conductor in the embodimentaccording to FIG. 3;

FIG. 5 shows the current and voltage distribution in an embodimentslightly altered relative to FIG. 3;

FIG. 6 shows a modification of the embodiment shown in FIG. 3;

FIGS. 7-9 show respective perspective views of third, fourth and fifthembodiments;

FIG. 10 shows a further modification of the embodiment according to FIG.3;

FIG. 11 shows a further development of the embodiment in FIG. 10; and

FIG. 12 shows a perspective view of a sixth embodiment of the couplingdevice according to the invention.

The coupling device shown in FIG. 1 comprises a waveguide section 1 ofrectangular cross-section, having an upper broad side wall 2, a lowerbroad side wall 3 and narrow side walls 8, in which the waveguide modeH₁₀ is capable of propagation. This waveguide mode has non-vanishingfield components H_(x), H_(z) and E_(y), where H_(x) and E_(y) areproportional to sin(πx/a) and H_(z) is proportional to cos(πx/a), wherea is the width of the broad side walls 2, 3 and the narrow side walls 8lie at coordinate values x=0 and x=a in the xyz-coordinate system shown.The field component H_(z) has a nodal plane at x=a/2.

A first slit 4 extends in the upper broad side wall 2 in the directionof the z axis. A second slit 5 is arranged relative to the nodal planex=a/2 as a mirror image of the first slit 4. Fields emerging from thetwo slits 4, 5 are composed of contributions from the non-vanishingfield components passing through the slit, and electric fields in thex-direction resulting from the fact that the slits 4, 5 block the pathof cross-currents flowing in the waveguide wall and evoked by thewaveguide mode. These cross-currents, illustrated schematically in FIG.2, have opposite signs on different sides of the nodal plane x=a/2. Thenodal plane is represented by chain-dashed lines M. Their contributionto the emerging fields is greater the stronger the cross-currents are atthe site of the slits 4, 5, i.e. the further these are removed from thenodal plane. The contributions of the cross-currents and of thecomponent H_(z) of the waveguide mode to the field outside the waveguidehave opposite signs on different sides of the nodal plane, so that thesefields cancel each other out in the radiation zone. The field componentsH_(x), E_(y) have the same sign on both sides of the nodal plane, sothat they do not cancel each other out in the radiation zone, althoughtheir field strength approaches zero with increasing proximity to thenarrow side walls 8, so that their contribution to the field outside thewaveguide section also is smaller the nearer the slits 4, 5 lie to thenarrow side walls 8.

On the upper broad side wall 2 is arranged a dielectric substrate 6,which bears a first strip line 7 bridging the first slit 4. The stripline 7 serves as an antenna section in which an electromagneticoscillation is induced by the electric field evoked by thecross-currents. This oscillation may be used to feed an antenna elementof a group antenna or another RF component.

A second strip line 9 may be arranged in mirror image fashion to thestrip line 7 over the second slit 5. Its function is the same as that ofthe first strip line; it may be used to feed the same RF component asthe first strip line 7, or a second RF component.

In the second embodiment of the coupling device according to theinvention shown in FIG. 3, the waveguide section 1 is the same as inFIG. 1 and will therefore not be described again. Two strip lines 7′, 9′formed on a substrate 6 extend on a common line parallel to the X-axisand are linked to each other at their ends facing each other and joinedto a common connecting conductor 10.

In the embodiment according to FIG. 3, the connecting point 11 of theends facing each other of the connecting conductor 10 lies on the nodalplane x=a/2 of the field component H_(z). It is to be noted that, inthis and subsequent figures, the lower of the two lines M delineatingthe plane x=a/2 shown in FIG. 2 has been omitted for clarity. Thespacing of the crossing points 12 of the strip lines 7′, 9′ from theirrespective free ends 13 is λ_(s)/4, and the spacing of the two crossingpoints 12 is λ_(s)/2, where λ_(s) is the wavelength of the oscillationinduced in the strip lines by the waveguide mode. The two strip lines7′, 9′ thus form a resonator matched to the waveguide mode of lengthλ_(s). In the resonator a standing wave forms, whose current and voltagepattern is illustrated by the dotted curve 1 and the dot-dashed curve Uin FIG. 4. At the connecting point 11, there is a node in the currentdistribution. The amplitude of the voltage is a maximum here, so that astrong signal may be drawn off via the connecting conductor 10.

In the variation shown in FIG. 5, the connecting point 11 does not liecentrally between the two free ends 13, but is displaced towards thefree end of the strip line 7′. The voltage level difference at theconnecting point 11 is lower than in the case in FIG. 4, and the signaldrawn off via the connecting conductor 10 is weaker. It is thereforepossible, independently of a coupling coefficient required in anindividual case, to manufacture the waveguide section 1 with the slits4, 5, the substrate 6 and the strip lines 7′, 9′ in a standard form andthrough contacting of the connecting conductor 10 at a suitably selectedconnecting point 11, to realise a coupling strength required in anindividual case.

Variable coupling coefficients are also realisable with the designaccording to FIG. 3 if, on the one hand, the waveguide section 1 and, onthe other hand, the substrate 6 with the strip lines 7′, 9′ situated onit and the connecting conductor 10 are manufactured in a standard form.In order to vary the coupling, it is sufficient to vary the position ofthe substrate and the conductors situated on it transverse to the nodalplane x=a/2. This leads to a deviation of the spacing between thecrossing points 12 and the free ends 13 from the optimum value λ_(s)/4.

By suitable selection of the position of the substrate 6, it is thuspossible to set the strength of the coupling between the waveguidesection 1 and the strip lines 7′, 9′. This significantly simplifies themanufacture of coupling devices with different coupling strengths, sinceit is not necessary to set the position of the slits 4, 5 according to adesired coupling strength and to manufacture a plurality of waveguidesections with differing slit spacings, but the waveguide sections 1 maybe manufactured in large quantities with a fixed position of the slitsand the desired coupling strength may be subsequently selected bysuitable positioning of the substrate 6.

Naturally, the spacings of the crossing points 12 from the free ends 13and the spacings of the crossing points 12 from each other do not haveto be λ_(s)/4 and λ_(s)/2, respectively, at the same time. Indeed,strong coupling may be achieved with such spacings, but only within avery narrow frequency range. If, for at least one of these spacings, anot exactly optimal value is chosen, but rather one lying close to it,then at somewhat reduced coupling strength, the bandwidth of thecoupling device may be significantly increased.

A variation of the principle in FIG. 3 is shown in FIG. 6. The waveguidesection 1 is the same again as in FIGS. 1 and 3, and the strip lines 7″,9″ deposited on the substrate 6 differ from those in FIG. 3 in that theresonator formed by them is C-shaped, and that the free ends 13 of theconductor sections 7″, 9″ both lie between the slits 4, 5. The method ofoperation otherwise corresponds to that of the example in FIG. 3.

The embodiment shown in FIG. 7 differs from that previously consideredin that in this case the two strip lines 7*, 9* formed on the substrate6 cross the slits 4, 5 of the waveguide section 1 assigned to them inthe same direction; their free ends 13 lie, respectively, on the side ofthe slits 4, 5 facing towards the viewer in the perspective of FIG. 7.For strong coupling of the strip lines 7*, 9* to the waveguide section1, a cophasal overlaying of the oscillations coupled into the two striplines 7*, 9* and thus a spacing between the two crossing points 12 ofthe slits 4, 5 with the strip lines 7*, 9* of (n+½)λ_(s) is required.The strength of the signal drawn off at the connecting conductor 10 maybe influenced, as in the example in FIG. 3, by selecting the position ofthe connecting points 11 of the connecting conductor 10 and by selectingthe spacing between the crossing points 12 and the free ends 13 of thestrip lines.

A particularly simple embodiment with strip lines 7**, 9** crossing theslits 4, 5 of the waveguide section 1 in the same direction is shown inFIG. 8. The strip line 9** crossing the slit 5 is connected in seriesbetween the strip line 7** and the connecting conductor 10. The crossingpoints 12 have a spacing from the single free end 13 of λ_(s)/4 and3λ_(s)/4, respectively.

FIG. 9 shows a further embodiment with strip lines 7***, 9*** connectedin series and crossing the slits 4, 5 in the same direction.

A further embodiment of the coupling device is shown in FIG. 10. Here,the substrate 6 and the strip lines 7′, 9′ formed thereon are identicalto those in FIG. 3; the waveguide section 1′ has been altered. Its slits4′, 5′ run parallel to each other, but at a non-vanishing angle α to thenodal plane x=a/2. Slit 4′ can be considered to the inverted reflectionof slit 5′ about the nodal plane.

The length of the slits in the Z-direction is chosen such that the phasedifference of the fields at opposing ends of the slits 4′, 5′ is notmore than 15°. The angle α influences the strength of the H_(z)component of the waveguide mode emerging through the slits 4′, 5′ andthus the strength of the magnetically induced current in the strip lines7′, 9′. At an angle α=0, this is a maximum; at a value of 90°, it wouldvanish.

A further development of this embodiment is shown in FIG. 11. Here, theslits 4′, 5′ are arranged in a circular disk 17 comprising part of theupper wall of the waveguide section 1′. Through rotation of the disk 17,the angle α between the slits 4′, 5′ and the nodal plane is variable andthe coupling strength may be adjusted.

FIG. 12 shows another embodiment of the coupling device in which thesubstrate 6 and the strip lines 7′, 9′ are identical to those in FIG. 3,while the waveguide section 1″, on the other hand, is modified. Itsslits 4″, 5″ run symmetrically to each other but inclined to the nodalplane x=a/2. The substrate 6 is displaceable in controlled mannerparallel to the nodal plane with the aid of laterally arranged guiderails 14, a micrometer screw 15 and a spring 16, in order thus toposition the strip lines 7′, 9′ over regions of the slits 4″, 5″ atdifferent spacings. As already stated in the explanation aboveconcerning the operation of the device, on displacement of the striplines 7′, 9′ the coupling varies, on the one hand, because the spacingof the crossing points 12 from each other and from the free ends 13changes and therefore the interference of the two signals induced in thetwo strip lines alters and, on the other hand, because the fields towhich the strip lines 7′, 9′ are exposed are all the stronger the nearerthe crossing points 12 lie to the side walls of the waveguide section1″. It is thus possible to set the coupling between the waveguidesection 1′ and the strip lines 7′, 9′ at any time precisely to acurrently-required value by displacing the substrate 6 along the Z-axis.

Naturally, with the embodiments in FIGS. 3 and 6 to 9, a rail guide maybe employed for controlled displacement of the substrate transverse tothe nodal plane x=a/2. Similarly, it is possible to permanently fix thesubstrate 6 on the waveguide section 1″ of FIG. 12 in a positionselected in beforehand according to a desired coupling strength, e.g. bycementing.

A plurality of the aforementioned coupling devices may be arranged alonga single waveguide. The spacing between the individual coupling devicesshould then be half the wavelength λ_(H) of the wave in the waveguide,so that the residual scattering fields of the individual couplingdevices cancel each other out in the radiation zone.

1-20. (canceled)
 21. A waveguide coupling device, comprising: awaveguide section in which a guided wave is propagated in at least onewaveguide mode, the waveguide section having a first slit in one of itswalls to form a slotted wall, the at least one waveguide mode having afield component parallel to the slotted wall with a nodal plane orientedin a longitudinal direction of the waveguide section and inducing in thewalls of the waveguide section a wall current distribution with justsaid nodal plane; a first antenna section bridging the first slit; and asecond slit in the slotted wall, the two slits lying on different sidesof the nodal plane.
 22. The coupling device according to claim 21, inthat the two slits are arranged in regions of opposing equal fieldstrength of the parallel field component.
 23. The coupling deviceaccording to claim 21, in that a locus of one of the two slits is areflection ofthe other of the two slits with respect to the nodal plane.24. The coupling device according to claim 21, in that a locus of one ofthe two slits is an inverted reflection of the other of the two slitswith respect to the nodal plane.
 25. The coupling device according toclaim 21, in that one free end of the first antenna section is placed ata spacing of λ_(s)/4 from the first slit, wherein λ_(s) is a wavelengthof an oscillation induced in the first antenna section by the guidedwave.
 26. The coupling device according to claim 21, in that thewaveguide section has a rectangular cross-section, and in that the atleast one waveguide mode is an H₁₀ mode.
 27. The coupling deviceaccording to claim 21, in that a second antenna section bridges thesecond slit.
 28. The coupling device according to claim 27, in that theantenna sections are linked at a point in parallel with a connectingconductor, in that the antenna sections cross the first and second slitsin respectively opposing directions, and in that the antenna sectionshave a total length L, wherein (n−⅜)λ_(s)<L<(n+⅜)λ_(s), wherein n is aninteger, and wherein λ_(s) is a wavelength of an oscillation induced inthe antenna sections by the guided wave.
 29. The coupling deviceaccording to claim 27, in that the antenna sections are joined at apoint in parallel with a connecting conductor, in that the antennasections cross the first and second slits in the same direction, and inthat the antenna sections have a total length L, wherein(n+⅛)λ_(s)<L<(n+⅞)λ_(s), wherein n is an integer, and wherein λ_(s) is awavelength of an oscillation induced in the antenna sections by theguided wave.
 30. The coupling device according to claim 27, in that theantenna sections are linked in series with a connecting conductor, inthat the antenna sections cross the first and second slits in opposingdirections, and in that a spacing between the slits measured along theantenna sections is between (n−⅜)λ_(s) and (n+⅜)λ_(s), wherein n is aninteger, and wherein λ_(s) is a wavelength of an oscillation induced inthe antenna sections by the guided wave.
 31. The coupling deviceaccording to claim 27, in that the antenna sections are linked in serieswith a connecting conductor, in that the antenna sections cross thefirst and second slits in the same direction, and in that a spacingbetween the slits measured along the antenna sections is between(n+⅛)λ_(s) and (n+⅞)λ_(s), wherein n is an integer, and wherein λ_(s) isa wavelength of an oscillation induced in the antenna sections by theguided wave.
 32. The coupling device according to claim 27, in that acrossing point of the antenna sections and the slits lies on a lineperpendicular to a longitudinal direction of the waveguide section. 33.The coupling device according to claim 27, in that the antenna sectionsare positioned at different positions transverse to the nodal plane. 34.The coupling device according to claim 21, in that the first and secondslits are parallel to each other and to the nodal plane.
 35. Thecoupling device according to claim 21, in that the first and secondslits run inclined to the nodal plane.
 36. The coupling device accordingto claim 35, in that the first and second slits are arranged in arotatable wall section of the waveguide section.
 37. The coupling deviceaccording to claim 27, in that the first and second slits have a spacingvarying along the nodal plane, and in that the antenna sections arepositionable at different positions along the nodal plane.
 38. Thecoupling device according to claim 27, and an apparatus for adjustingthe antenna sections relative to the slits.
 39. The coupling deviceaccording to claim 27, in that the antenna sections are strip linesections arranged on a substrate.
 40. The coupling device according toclaim 21, in that further slits are formed in the slotted wall of thewaveguide section at a spacing of (n+½)λ_(H), wherein λ_(H) is awavelength of the guided wave in the waveguide section.